Wireless Full-Duplex System and Method Using Sideband Test Signals

ABSTRACT

A full-duplex transceiver is provided with componentry and methods for cancellation of nonlinear self-interference signals. The transceiver is capable of receiving an incoming radio-frequency signal that includes both a desired radio-frequency signal component and a self-interference component caused by the transceiver&#39;s own radio-frequency transmission. The transceiver demodulates the incoming radio-frequency signal to generate a first demodulated signal. The transceiver combines an analog corrective signal with the first demodulated signal to generate a second demodulated signal with reduced self-interference. The transceiver processes the first and second demodulated signals to determine a desired incoming baseband signal and to determine nonlinear components of the self-interference signal, such as nonlinearities introduced by the transceiver&#39;s power amplifier.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application a continuation of U.S. application Ser. No. 15/334,013,filed Oct. 25, 2016, titled “WIRELESS FULL-DUPLEX SYSTEM AND METHODUSING SIDEBAND TEST SIGNALS”, which is a continuation of U.S.application Ser. No. 14/993,797, filed Jan. 12, 2016, now U.S. Pat. No.9,479,322, which is a continuation of U.S. application Ser. No.14/301,088, filed Jun. 10, 2014, now U.S. Pat. No. 9,236,996, whichclaims the benefit under 35 U.S.C. § 119(e) from the followingapplications: U.S. Provisional Patent Application Ser. No. 61/910,332,filed Nov. 30, 2013, and U.S. Provisional Patent Application Ser. No.61/916,511, filed Dec. 16, 2013, all of which are incorporated herein byreference in their entirety.

FIELD

The present disclosure relates to wireless communications. Inparticular, the present disclosure relates to systems and methods toestablish two-way (full-duplex) wireless links.

BACKGROUND

A communication link with capability to support connections in bothtransmit and receive directions at the same time and over the entirefrequency band is called full-duplex, or two-way. In contrast, a linkthat can support connection in only one direction at a time (over agiven frequency band) is called one-way or half-duplex. Current wirelesssystems are one-way and rely on either separate time slots (timedivision duplex) or separate frequency bands (frequency division duplex)to transmit and to receive. These alternatives have their pros and cons,but both suffer from lack of ability to transmit and to receiveconcurrently over entire frequency band. Even in the context ofOrthogonal Frequency Division Multiple Access (OFDMA), where differentfrequency tones are used to simultaneously service multiple users, thereis no method known to use the OFDM tones in opposite directions. Asimilar shortcoming exists in the context of Code Division MultipleAccess (CDMA). Although full-duplex wireless is theoretically possible,its implementation is difficult due to an excessive amount ofinterference caused by a transmitter to its own receiver(s).

Full-duplex communication is currently used in many applications, e.g.,wired telephones, digital subscriber line, wireless with directionalantennas, and free-space optics. The impact of full-duplex links inthese earlier applications is limited to doubling the communicationsrate by providing two symmetrical pipes of data flowing in oppositedirections. In contrast, in multi-user wireless systems, due to thebroadcast nature of transmission (everyone hears everyone else),full-duplex capability has the potential to do more than merely doublethe communications rate. A summary of some of the benefits offull-duplex is as follows.

Full-duplex facilitates wireless networking. In particular, the abilityto handle asynchronous users enables superimposing a half-duplex, lowbit rate, low power, easy to detect network for control signalingsuperimposed (physical overlay, rather than logical) on top of thenetwork of primary full-duplex data links. The superimposed links areseparated from the primary full-duplex data links in the code domain,and use time multiplexing plus Carrier Sense Multiple Access (CSMA)among themselves. However, the conventional problems of CSMA are avoidedas control links operate in parallel with primary full-duplex datalinks. The physical layer of control links is designed such thatfull-duplex data links can detect and cancel the interference caused bythe superimposed control links.

Full-duplex enhances security through desirable jamming.

Full-duplex facilitates multi-node distributed & collaborativesignaling, including realization of Network Information Theoreticsetups, and cognitive wireless.

Full-duplex, through exploiting feedback, improves point-to-pointthroughput, and enables ultra-low power transmission.

Full-duplex doubles the point-to-point throughput.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

The accompanying figures, where like reference numerals refer toidentical or functionally similar elements throughout the separateviews, together with the detailed description below, are incorporated inand form part of the specification, and serve to further illustrateembodiments of concepts that include the claimed invention, and explainvarious principles and advantages of those embodiments.

FIG. 1 is a functional block diagram of a full-duplex transceiveraccording to an embodiment described herein.

FIG. 2 is a flow diagram of a process performed in a full-duplextransceiver according to an embodiment described herein.

FIG. 3 is a flow diagram of another process performed in a full-duplextransceiver according to an embodiment described herein.

FIG. 4 is a schematic graph illustrating the spectral power density ofan outgoing communication signal combined with a pair of sideband testsignals according to an embodiment described herein.

FIG. 5 is another schematic graph illustrating the spectral powerdensity of an outgoing communication signal combined with a pair ofsideband test signals according to an embodiment described herein.

FIG. 6 is a functional block diagram of a receive chain used to measuresignal distortion in an embodiment described herein.

FIG. 7 is a functional block diagram of a receive chain used to measuresignal distortion in another embodiment described herein.

FIG. 8 is a functional diagram of receive circuitry for use with anintermediate-frequency (IF) cancellation signal.

FIG. 9 is a functional block diagrams illustrating componentry forgenerating a baseband and a radio-frequency corrective signal.

FIG. 10 is a functional block diagrams illustrating componentry forgenerating an intermediate-frequency and a radio-frequency correctivesignal.

FIG. 11 is a schematic block diagram of a full-duplex transceiver usingtwo receive chains.

FIGS. 12 to 16 show embodiments for a dual mode system operating ineither a full-duplex mode, or in a MIMO mode.

FIG. 12 shows the MIMO mode as a transmitter.

FIG. 13 shows the MIMO mode as a receiver.

FIG. 14 shows an embodiment of a dual mode system, wherein the hardwarepieces in transmit and receive chains of MIMO modes are reused infull-duplex mode.

FIG. 15 shows another embodiment of a dual mode system, wherein thehardware pieces in transmit and receive chains of MIMO modes are reusedin full-duplex mode.

FIG. 16 shows another embodiment of a dual mode system, wherein thehardware pieces in transmit and receive chains of MIMO modes are reusedin full-duplex mode.

FIG. 17 shows an embodiment using an auxiliary transmit signal foranalog active cancellation, and an auxiliary receive signal usedsampling of the PA output, wherein the hardware complexity is reduced byusing a multiple terminal antenna.

FIG. 18 shows an embodiment using an auxiliary corrective signal foranalog active cancellation in analog base-band, and an auxiliarybase-band receive signal used for the sampling of the PA output tocapture PA non-linearity effect as well as PA noise.

FIGS. 19 to 22 show embodiments using two auxiliary corrective signals,one for analog active cancellation in RF, and one for analog activecancellation in either base-band or IF, and an auxiliary receive signalis used for sampling of the PA output to capture PA non-linearity effectas well as PA noise.

FIG. 23 shows a high level pictorial view for the use of AF relays.

FIG. 24 shows a high level pictorial view for an AF relay with analog RFself-interference cancellation.

FIG. 25 shows a high level pictorial view for an AF/DF relay with analogRF self-interference cancellation, wherein the decision about the AF/DFmode of operation is performed during OFDM signal preamble such that theAF mode can be selected fast enough to remain transparent to the rest ofthe network.

FIG. 26 shows a hardware simplification by directly filtering the signalin the primary transmit chain to produce the corrective signal.

DETAILED DESCRIPTION OF THE INVENTION

The present disclosure describes systems and methods performed in afull-duplex transceiver. Techniques disclosed in the context offull-duplex wireless can be applied to cable modem, Ethernet Cable, aswell as to fiber optical channels.

In one such method, the transceiver receives an incoming radio-frequencysignal that includes both a desired radio-frequency signal component anda self-interference component caused by the transceiver's ownradio-frequency transmission. The transceiver demodulates the incomingradio-frequency signal to generate a first demodulated signal. Thetransceiver also generates a first analog corrective signal and combinesthe first analog corrective signal with the first demodulated signal togenerate a second demodulated signal with reduced self-interference. Thetransceiver samples the first demodulated signal to obtain a sampledself-interference component; and the transceiver processes at least thesecond demodulated signal and the sampled self-interference component todetermine a desired incoming baseband signal. The transceiver may alsogenerate a second analog corrective signal at a radio frequency that iscombined with the incoming radio-frequency signal before demodulation tocancel at least a portion of the self-interference component.

The generation of the first and second corrective signals may beperformed by generating a combined corrective signal at adigital-to-analog converter. The combined corrective signal is modulatedto generate the second analog corrective signal at the radio frequency.The combined corrective signal is also filtered to generate the firstanalog corrective signal. In embodiments in which the combinedcorrective signal includes a baseband corrective signal component, thecombined corrective signal is low-pass filtered to generate a basebandcorrective signal. In embodiments in which the combined correctivesignal includes an intermediate-frequency corrective signal component,the combined corrective signal is band-pass filtered to generate anintermediate-frequency corrective signal.

In order to recover the data from the desired incoming baseband signal,the transceiver may perform maximal-ratio combining of at least thesecond demodulated signal and the sampled self-interference component.Alternatively, or in addition, the transceiver may perform minimum meansquare error processing of at least the second demodulated signal andthe sampled self-interference component.

During this signal cancellation process, the transceiver may alsosimultaneously transmit outgoing radio-frequency signals onsubstantially the same frequency as the incoming radio-frequency signal.To do this, the transceiver may modulate an outgoing baseband signal togenerate an outgoing radio-frequency communication signal. Thetransceiver then amplifies the outgoing radio-frequency communicationsignal using a power amplifier and transmits the outgoingradio-frequency communication signal from a transmit antenna. It is thissimultaneous transmission that generates the self-interference componentreceived at the receive antenna of the full-duplex transceiver.

To generate the first analog corrective signal, the transceiver mayapply a predetermined transformation to a local copy or replica of theoutgoing baseband signal. In addition, the transceiver may process thesampled self-interference component to determine the nonlineardistortion introduced by the power amplifier, and this informationregarding the nonlinear distortion may be used in determining thepredetermined transformation to be applied to the local copy of theoutgoing baseband signal to generate the first analog corrective signal.Thus, in this embodiment, the nonlinear distortion caused by the poweramplifier is measured and modeled based on the sampled self-interferencecomponent, and the model is used to create the first analog correctivesignal.

In some embodiments, the first analog corrective signal and the firstdemodulated signal are intermediate-frequency (IF) signals. Thedemodulation of the incoming radio-frequency signal to generate thefirst demodulated signal may be performed without the use ofimage-rejection circuitry.

In another method performed by a full-duplex transceiver, a combinedradio-frequency communication signal is generated. The combined signalincludes (i) an outgoing radio-frequency communication signal that hasone center frequency (i.e., a band of frequencies centered at one centerfrequency), and (ii) a radio-frequency test signal that has a differentcenter frequency. The transceiver amplifies the combined radio-frequencycommunication signal using a power amplifier and transmits the amplifiedcombined radio-frequency communication signal from a transmit antenna.Due to self-interference, the full-duplex transceiver also receives anincoming radio-frequency signal that includes a radio-frequency testcomponent caused by the transmitted radio-frequency test signal and aself-interference component caused by the transmitted radio-frequencycommunication signal. The transceiver demodulates the incoming signal,and samples the demodulated test component to obtain a sampled testcomponent. The transceiver then processes the sampled test component todetermine nonlinear distortion introduced by the power amplifier. Thetransceiver may also generate a radio-frequency corrective signal thatis combined with the incoming radio-frequency signal to cancel at leasta portion of the self-interference component. Preferably, theradio-frequency corrective signal cancels a substantial portion of theself-interference component but does not cancel the radio-frequency testcomponent.

The sideband radio frequency test signal has a power substantially lessthan the power of the radio frequency communication signal, preferably apower at least 20 dB less than the power of the radio frequencycommunication signal. Preferably, the sideband radio frequency testsignal has a power substantially greater than the power of the desiredincoming radio-frequency signal component. For example, the sidebandradio test signal may have a power at least 20 dB greater than the powerof the desired incoming radio-frequency signal component.

The combined radio-frequency communication signal may further include asecond radio-frequency test signal, where the first radio-frequency testsignal has a center frequency lower than the center frequency of theoutgoing radio-frequency communication signal, and the second radiofrequency test signal has a center frequency higher than the centerfrequency of the outgoing radio-frequency communication signal. In thiscase, the incoming radio-frequency signal includes a first and a secondradio-frequency test component caused by the respective first and secondradio-frequency test signals. The demodulation thus involvesdemodulating both the first and second radio-frequency test componentsto obtain respective first and second demodulated test components.Likewise, the sampling process then includes sampling the first andsecond demodulated test component to obtain respective first and secondsampled test components. The transceiver then processes both the firstand second sampled test components to determine nonlinear distortionintroduced by the power amplifier.

The radio-frequency test signal may include a plurality of subcarriersignals. In this case, the transceiver can processes the subcarriersignals to determine intermodulation distortion between the subcarriersignals.

A full-duplex transceiver described herein includes a radio-frequencymodulator operative to modulate an outgoing baseband signal into anoutgoing radio-frequency communication signal. The transceiver furtherincludes a transmit antenna operative to transmit the outgoingradio-frequency communication signal and a receive antenna operative toreceive an incoming radio-frequency signal. The transceiver includes aradio-frequency demodulator, which is operative to demodulate theincoming radio-frequency signal to generate a first demodulated signal.The transceiver further includes a baseband correction module that isoperative to generate a baseband analog corrective signal based on theoutgoing baseband signal, as well as baseband signal addition circuitry,which adds the baseband analog corrective signal to the firstdemodulated signal to generate a second demodulated signal with reducedself-interference. A first analog-to-digital converter works to samplethe first demodulated signal, and a second analog-to-digital converterworks to sample the second demodulated signal. A digital decoding moduleof the transceiver is operative to determine a desired incoming basebandsignal based at least on the sampled first and second demodulatedsignals. The transceiver may also include a radio-frequency correctionmodule that generates a radio-frequency analog corrective signal, andradio-frequency signal addition circuitry that adds the radio-frequencyanalog corrective signal to the incoming radio-frequency signal to aidin cancelling the self-interference signal.

As illustrated in FIG. 1, a full duplex transceiver 100 is provided withboth a transmit antenna 102 and a receive antenna 104. The full duplextransceiver 100 is preferably operable to transmit radio frequencysignals on the transmit antenna 102 while simultaneously receiving radiofrequency signals on the receive antenna 104 on the same frequency.

To transmit signals, the transceiver 100 includes a baseband signalgenerator 106 that generates the outgoing baseband signals containinginformation to be transmitted in digital form. In some embodiments, thebaseband signal generator 106 generates orthogonal frequency divisionmultiplexing (OFDM) signals, which employ quadrature amplitudemodulation (QAM) and thus have an in-phase component (I) and aquadrature component (Q). The baseband signal may encode any of avariety of types of information, including voice and data information,according to any of a variety of coding schemes known to those skilledin the art. While the use of OFDM baseband signals is illustrated hereby way of example, the principles described herein can be implementedusing alternative types of signaling, such as phase-shift keying (PSK),frequency-shift keying (FSK), or amplitude-shift keying (ASK), amongothers.

The digital in-phase (I) and quadrature (Q) components of the basebandsignal are converted to analog baseband signals using respectivedigital-to-analog converters 108, 110. In some embodiments, the signalsprovided by the baseband signal generator 106 are digital signals in thefrequency domain. In such embodiments, the digital signals in thefrequency domain may be converted into digital signals in the timedomain by application of an inverse fast Fourier transform (inverseFFT), for example by a dedicated inverse FFT chip (not illustrated),prior to conversion to analog signals in the time domain by thedigital-to-analog converters 108, 110.

A local oscillator 112 provides a radio-frequency signal that is used bya radio-frequency modulator 114 to modulate the analog basebandcommunication signal to radio frequency. The radio-frequency modulator114 may include a frequency mixer. In embodiments that make use of OFDMsignaling, the in-phase and quadrature analog components of the analogbaseband signal may be separately modulated by respective in-phase andquadrature components of the local oscillator signal and may then beadded together for transmission.

The radio-frequency communication signal generated by theradio-frequency modulator 114 is then amplified by a power amplifier116. While an ideal power amplifier would simply increase the amplitudeof the radio-frequency communication signal by a scalar factor, inpractice, the power amplifier 116 introduces nonlinear distortions inthe signal being amplified. The power amplifier 116 is coupled to thetransmit antenna 102 to wirelessly transmit the radio-frequencycommunication signal.

To permit simultaneous transmission and receipt of radio frequencysignals by the full-duplex transceiver 100, it is desirable to minimizethe interference at the receive antenna 104 caused by the transmitantenna 102. Nonlinear compensation methods will generally not providesatisfactory results if the level of nonlinear leakage is high. Due tothis reason, it is important to minimize the coupling between transmitand receive chains prior to active cancellation. This enables a level ofoverall isolation that would not be feasible through linear signalprocessing techniques. As an example, numerical errors in a 14 bit,64-point inverse FFT operation, with an optimized fixed-point arithmeticdesign, are about −80 dB below the signal level. This is also inaccordance with the theoretical resolution of a 14 bit digital-to-analog(D/A) converter, which is widely available and a reasonable D/A choicefor cost effective implementation. Under this condition, a transmitsignal at 30 dBm can result in numerical errors at −60 dBm. Ifradio-frequency isolation between transmit and receive chain is −50 dB,then the numerical errors (at −110 dBm) will be below the thermal noiselevel. In practice, such errors should be comparable to the noise level,which enables subsequent signal processing to account for and compensatethe nonlinear effects.

One way to help minimize the interference at the receive antenna 104caused by the transmit antenna 102 is to design and orient the transmitantenna 102 and the receive antenna 104 to minimize electromagneticcoupling between those antennas. For example, the transmit and receiveantennas may be pairwise symmetrical. That is, the transmit and receiveantennas have separate planes of symmetry, and each antenna is alsosymmetric with respect to reflection in the plane of symmetry of theother antenna. As an alternative, patch antennas can be used in whichone arm of an antenna is generated through reflection of the other armin the ground plane. As another alternative to pairwise symmetricalstructures, it is possible to place one set of antennas in the plane ofsymmetry of another set, which is shown to be an equipotential surface.While such orientations theoretically reduces coupling between theantennas to zero, in practice, some coupling still remains between thetransmit antenna 102 and the receive antenna 104.

During operation of the full-duplex transceiver 100, the receive antenna104 receives an incoming radio-frequency signal. The incomingradio-frequency signal can include both a desired radio-frequency signalcomponent (transmitted by another transmitter) and a self-interferencecomponent caused by transmission from the transmit antenna 102.

To cancel out at least a portion of the self-interference component, thefull duplex transceiver 100 is provided with radio frequency signaladdition circuitry 118 to combine the incoming radio-frequency signalwith an analog radio-frequency corrective signal. To generate the analogradio-frequency corrective signal, the transceiver 100 is provided witha radio frequency correction module 120. The radio frequency correctionmodule 120 processes the digital baseband signal from the basebandsignal generator 106 to generate a digital baseband signal that isconverted by analog-to-digital converters 122, 124 into an analogbaseband signal. This analog baseband signal is in turn modulated by aradio-frequency modulator 126 to generate the analog radio-frequencycorrective signal. The radio-frequency modulator 126 preferably makesuse of the same local oscillator 112 as the radio-frequency modulator114.

The radio-frequency correction module 120 is configured to process theoutgoing baseband signal such that the analog radio-frequency correctivesignal generated at the radio-frequency modulator 126 cancels at least aportion of the self-interference component when the signals are combinedat the radio-frequency signal addition circuitry 118. For example, theradio-frequency correction module 120 may apply a transfer function tothe baseband signal that imitates the coupling characteristics along thepath from the baseband signal generator 106, through the transmitantenna 102, to the receive antenna 104. The radio-frequency correctionmodule 120 thus increases the extent to which the analog radio-frequencycorrective signal cancels the self-interference component at theradio-frequency signal addition circuitry 118. The radio-frequencysignal addition circuitry 118 may be, for example, a passive transformercircuit, or an RF coupler.

Preferably, after at least a portion of the self-interference signal hasbeen canceled at the radio-frequency signal addition circuitry 118, theresulting incoming radio-frequency signal is sufficiently low that itcan be fed to the input of a low-noise amplifier 128 without saturatingthe input. The radio-frequency signal is amplified by the low-noiseamplifier 128 and is demodulated by a radio-frequency demodulator 130.The radio-frequency demodulator is preferably coupled to the localoscillator 112. In the exemplary transceiver 100, the radio-frequencydemodulator 130 provides a demodulated signal that includes separateanalog in-phase (I) and quadrature (Q) components.

This demodulated signal contains self-interference components that maynot have been canceled at the radio-frequency addition circuitry 118. Toprovide additional self-interference cancellation, the transceiver 100is provided with baseband signal addition circuitry 134, 136, to combinethe demodulated signal with a baseband analog corrective signal. Thebaseband analog corrective signal is provided with a baseband correctionmodule 132. The baseband correction module 132 processes the digitalbaseband signal from the baseband signal generator 106 to generate adigital baseband signal that is converted by analog-to-digitalconverters 138, 140 into a baseband analog corrective signal.

The baseband correction module 132 is configured to process the outgoingbaseband signal such that the baseband analog corrective signal cancelsat least a portion of the self-interference component when the signalsare combined at the baseband signal addition circuitry 134, 136. Forexample, when the baseband signal is stored as digital values in thefrequency domain, the baseband correction module 132 may apply atransfer function to the baseband signal that imitates the couplingcharacteristics along the path from the baseband signal generator 106,through the transmit antenna 102, to the receive antenna 104, andthrough the demodulator 130. In some embodiments, the basebandcorrection module 132 adjusts the amplitude of the baseband analogcorrective signal. The baseband correction module 132 thus increases theextent to which the baseband analog corrective signal cancels theself-interference component at the baseband signal addition circuitry134, 136. The baseband signal addition circuitry 134, 136, may be, forexample, a pair of passive transformer circuits, it may be a pair ofop-amp signal adders, or it may take other forms.

After the baseband signal passes through the signal addition circuitry134, 136, the I and Q components of the baseband signal are sampled byrespective analog-to-digital converters 142, 144 and converted todigital form for further processing. In parallel, anotheranalog-to-digital converter 146 samples the demodulated signal that hasnot passed through the baseband signal addition circuitry 134, 136. Theanalog-to-digital converter 146 may consist of a pair ofanalog-to-digital converters (analogous to 142, 144) that separatelydigitize the I and Q components of the baseband signal, oranalog-to-digital converter 146 may be a single analog-to-digitalconverter that takes alternate samples of the I and Q components of thebaseband signal.

As a consequence of the self-interference cancellation at the signaladdition circuitry 134, 136, the signal sampled by analog-to-digitalconverters 142, 144 has a greatly reduced self-interference component,and consequently, the component of the signal due to the desiredincoming signal is relatively stronger. On the other hand, the signalsampled at the analog-to-digital converter (or converters) 146 has notexperienced baseband signal cancellation, and as a result, it isdominated by the self-interference component. The signal sampled byanalog-to-digital converter 146 thus provides a measure of theself-interference component caused by radio-frequency transmissions fromthe transmit antenna 102.

The transceiver 100 is provided with a digital decoding module 148. Thedigital decoding module 148 is operative to process the signals sampledby the analog-to-digital converters 142, 144, 146 to determine thedesired incoming baseband signal. For example, the digital decodingmodule 148 may employ maximal-ratio combining of the demodulated signalfrom analog-to-digital converters 142, 144 with the sampledself-interference component sampled by the analog-to-digital converter146. As an alternative (or in combination), the digital decoding module148 may perform minimum mean square error processing of the demodulatedsignal from analog-to-digital converters 142, 144 and the sampledself-interference component sampled by the analog-to-digital converter146. Where the incoming baseband signal is an OFDM signal, the digitaldecoding module may operate by performing a fast Fourier transform (FFT)on the incoming signals to identify the subcarrier components of theOFDM signal. In such an embodiment, the digital decoding module 148 mayinclude a dedicated FFT chip.

The digital decoding module 148 may also operate to determine thenonlinear distortion introduced by the power amplifier 116. For example,the digital decoding module 148 may compare the undistorted outgoingbaseband signal from baseband signal generator 106 with the distortedself-interference component sampled by the analog-to-digital converter146. The digital decoding module 148 may generate a model of thedistortion introduced by the power amplifier 116, for example bytreating the distortion as the output of a Volterra series to which theinput is the original outgoing baseband signal. Based on theself-interference component sampled by the analog-to-digital converter146, the digital decoding module 148 may update the operation of theradio-frequency correction module 120 and the baseband correction module132 to improve the self-interference signal cancellation introduced bythose modules. For example, based on the nonlinear model (e.g., aVolterra series) of distortion introduced by the power amplifier 116,the RF correction module 120 may apply that same model to the outgoingRF corrective signal by modifying the corresponding baseband signal(prior to up-conversion) according to the measured nonlinearity. In thatway, the RF corrective signal more closely mimics the self-interferencecomponent and thus more fully cancels the self-interference component.Similarly, based on the nonlinear model (e.g., a Volterra series) ofdistortion introduced by the power amplifier 116, the basebandcorrection module 132 may apply that same model to the outgoing basebandsignal when the baseband analog corrective signal is generated. In thatway, the base-band corrective signal more closely mimics theself-interference component and thus more fully cancels theself-interference component.

The process of determining kernel coefficients of a Volterra seriesbased on empirical inputs is known to those skilled in the art. Forexample, the coefficients of an orthogonalized series such as a Wienerseries can be estimated, followed by computation of the kernelcoefficients of the Volterra series. Other nonlinear modeling techniquesmay alternatively be used.

As illustrated in FIG. 2, in a method performed at a full-duplextransceiver, an outgoing baseband signal is generated at step 202. Basedon the outgoing baseband signal, an analog radio-frequency correctivesignal is generated in step 204, and an analog baseband correctivesignal is generated in step 206. These corrective signals may begenerated by applying respective predetermined transformations to theoutgoing baseband signal. The predetermined transformations may take oneor more of several different forms. For example, the predeterminedtransformations may be implemented by applying a linear transferfunction to a digital baseband signal in the frequency domain, byapplying a filters to the baseband signal in the time domain, or byapplying a nonlinear transformation such as a Taylor series or aVolterra series.

In step 214, the transceiver receives an incoming radio-frequencysignal. The incoming radio frequency signal includes a desiredradio-frequency signal component sent by a remote transmitter. Infull-duplex operation, the transceiver is transmitting an outgoingradio-frequency signal at the same time and on the same frequency as theincoming radio-frequency signal. As a result, the incomingradio-frequency signal also includes a self-interference component.

In step 216, the transceiver combines the radio-frequency correctivesignal with the incoming radio-frequency signal to cancel out at least aportion of the self-interference component. In step 218, the resultingsignal is amplified by a low-noise amplifier. Preferably, thecombination of the radio-frequency corrective signal with the incomingradio-frequency signal in step 216 lowers the level of self-interferencesufficiently to prevent saturation of the input of the low noiseamplifier.

In step 220, the incoming radio frequency signal is demodulated togenerate a first demodulated signal. The first demodulated signal issampled in step 222, for example by one or more analog-to-digitalconverters, to obtain a sampled self-interference component. Inembodiments in which the first demodulated signal includes an in-phasecomponent (I) and a quadrature component (Q) step 222 may be performedby separately sampling the in-phase and quadrature components of thefirst demodulated signal with the use of two different analog-to-digitalconverters. As an alternative, step 222 may be performed by a singleanalog-to-digital converter that alternately samples the I component andthe Q component of the first demodulated signal. In this latter case,the rate of sampling may be increased by a factor of two as comparedwith the use of separate analog-to-digital converters.

In step 224, the sampled self-interference component is processed todetermine the nonlinear distortion in the self-interference component.For example, in step 220, the transceiver may compare the undistortedoutgoing baseband signal generated in step 202 with the distortedself-interference component sampled in step 222 to generate a model ofthe distortion. The distortion being modeled may include distortionintroduced by a power amplifier, low-noise amplifier, or othercomponents in the transmit and receive chain of the transceiver. Thenonlinear distortion may be modeled by, for example by treating thedistortion as the output of a Volterra series to which the input is theoriginal outgoing baseband signal.

In step 208, the first demodulated signal is combined with the basebandcorrective signal to generate a second demodulated signal. The basebandcorrective signal substantially cancels the self-interference componentof the incoming signal, but it does not cancel a component of theincoming signal that is attributable to a desired incoming signal from aremote transmitter.

In steps 216 and 208, the signals may be combined with the use of, forexample, a passive transformer circuit, a coupler, an op-amp signaladder, or other signal combination circuitry. As will be understood tothose of ordinary skill in the art, the signals being combined in steps216 and 208 are combined so as to effect signal cancellation. To dothis, the signal combination circuitry may be configured to subtract onesignal from another, or the signals being canceled may be phase-shiftedby 180° with respect to one another, or one of the signals may beinverted. Various schemes for canceling signals are known to thoseskilled in the art.

The generation of the corrective signals in steps 204 and 206 may beperformed based on the nonlinear distortion determined in steps 224 toimprove the cancellation of the self-interference component in steps 216and 208. For example, based on the nonlinear model (e.g., a Volterraseries) of distortion introduced by a power amplifier and/or othernonlinear components, the corrective signals generated in steps 204 and206 may be generated by applying that same model to the outgoingbaseband signal. In that way, the corrective signals more closely mimicthe self-interference component and thus more fully cancel theself-interference component.

In step 210, the second demodulated signal sampled by, for example, ananalog-to-digital converter. In step 212, the sampled signals areprocessed to determine an incoming baseband signal. For example, thetransceiver may employ maximal-ratio combining of the sampled firstdemodulated signal from step 222 with the sampled second demodulatedsignal from step 210. As an alternative (or in combination), thetransceiver may perform minimum mean square error processing of thesampled first demodulated signal from step 222 and the sampled seconddemodulated signal from step 210. Where the desired incoming basebandsignal is an OFDM signal, step 212 may involve performance of a fastFourier transform to obtain the OFDM subcarrier signals of the incomingOFDM signal.

In an alternative embodiment, an intermediate-frequency correctivesignal is used in place of, or in addition to, a baseband correctivesignal. For example, with respect to the method illustrated in FIG. 2,the generation of a baseband corrective signal in step 206 may bereplaced with the generation of an intermediate-frequency (IF)corrective signal. In such an embodiment, the first demodulated signalgenerated in step 220 may be an intermediate-frequency demodulatedsignal. The use of intermediate-frequency processing is useful infull-duplex communications to avoid the problems with the transmissionand consequently self-cancellation of DC components, as well as toreduce the effect of 1/f noise. In embodiments using intermediatefrequencies, it is desirable to have a LO-IF to RF modulator fortransmission, and an RF to LO-IF demodulator for reception. In such acase, the LO-IF to RF modulator as well as the RF to LO-IF demodulatorare preferably equipped with circuitry for image rejection. Anotherbenefit of working at LO-IF (instead of baseband) is that thecomplexities of constructing a corrective signal and of sampling a poweramplifier output can be reduced.

It should be noted that in RF to LO-IF demodulation of the poweramplifier output, there is no strong image present, and consequently, socomplicated image-rejection circuitry is not required, and the receiverfor sampling power amplifier output can be simplified. FIG. 8 shows anembodiment of receiver circuitry 800 for RF to LO-IF demodulation,wherein the circuit for image rejection is combined with the circuitneeded to sample the power amplifier output. In this case, the path forsampling the power amplifier output relies on direct sampling of the IFsignal with a single analog-to-digital converter 802. Subsequentdown-conversion to base-band and I/Q extractions of the PA sample areachieved using digital processing. Note that methods known for singleside-band modulation and Hilbert transform can be used to remove theimage in LO-IF to RF modulation.

Another benefit of working at LO-IF is that the RF corrective signal andthe lower frequency (intermediate frequency or baseband) correctivesignal can be constructed using a single chain by multiplexing these twocorrective signals in the frequency domain. FIGS. 9 and 10 illustrateembodiments wherein the construction of the two corrective signals isperformed with the use of a single transmission chain. As illustrated inFIG. 9, a baseband signal generator 902 is provided. From the basebandsignal, an intermediate frequency correction module 904 generates adigital intermediate frequency corrective signal, and a basebandcorrection module 906 generates a digital baseband corrective signal.These signals are multiplexed at a digital-to-analog converter 908,which generates a single combined analog signal with components in thebaseband and the intermediate frequencies. The combined analog signal ismodulated to radio frequency by an IF-to-RF modulator 910 to generate aradio frequency corrective signal. The combined analog signal isfiltered by a low-pass filter 912 to filter out the intermediatefrequency component, leaving the baseband corrective signal.

As illustrated in FIG. 10, a baseband signal generator 1002 is provided.From the baseband signal, an intermediate frequency correction module1004 generates a digital intermediate frequency corrective signal, and abaseband correction module 1006 generates a digital baseband correctivesignal. These signals are multiplexed at a digital-to-analog converter1008, which generates a single combined analog signal with components inthe baseband and the intermediate frequencies. The combined analogsignal is modulated to radio frequency by a baseband-to-RF modulator1010 to generate a radio frequency corrective signal. The combinedanalog signal is filtered by a band-pass filter 1012 to filter out thebaseband frequency component, leaving the intermediate frequencycorrective signal.

In the embodiments employing intermediate frequencies, differentcarriers are used for RF up-conversion and for RF down-conversion, andideally these carriers should have a fixed and stable difference intheir frequencies. To provide such stable frequency differences, in someembodiments, the mismatch is measured and accounted for(pre-compensated) in the construction of the low frequency correctivesignal. In other embodiments, one RF carrier is generated and mixed witha low frequency sinusoid to generate the second RF carrier, and the lowfrequency sinusoid is used as the reference in frequency multiplexing ofthe two corrective signals.

In some embodiments of the full-duplex transceiver, for situations inwhich the RF corrective signal is generated by re-modulating the primaryRF transmit signal, the phase and magnitude of the re-modulating carrieris adjusted to enhance the cancellation. In another embodiment, using acomponent that introduces an adjustable phase shift and an adjustablemagnitude scaling, the power amplifier output can be sampled directly inthe transmit chain, passing the corresponding RF signal through the unitto adjust its phase and magnitude, and injecting the resulting RFcorrective signal in the receive chain.

In another embodiment, multiple RF corrective paths are deployed inparallel, each coupled to the receive chain, wherein the correspondingcouplers to the receive chain are isolated using switchable (lowgain/high gain) LNAs. The phase and magnitude of the corrective pathsare adjusted sequentially. This means adjustment of the chain indexed byX+1 is performed after the phase and magnitude of the RF correctivesignal in earlier RF corrective chains, i.e., 1, . . . , X (lower indexis closer to the RX antenna) are first adjusted and fixed. Then, inadjusting chain X+1, the LNA separating the couplers corresponding tochains X and X+1 is turned to high gain, and then the phase andmagnitude of the corrective path is adjusted to minimize the residualenergy of the self-interference. Another embodiment of this inventionincludes incorporating a variable delay element in each RF correctivepath, which adjusts the delay in its corresponding RF corrective path tomatch the delay of the leakage path it aims to cancel.

In such embodiments, a method for providing self-interferencecompensation includes sampling the outgoing radio-frequency signal atthe radio frequency transmission chain. Preferably, the outgoingradio-frequency signal is sampled at the output of the power amplifier.The sampling may be performed with the use of a radio frequency powerdivider. The phase and magnitude of the sampled radio-frequency signalis adjusted to generate a radio-frequency corrective signal. Theadjustment may be performed with the use of a delay line. The delayintroduced by the delay line preferably corresponds to the travel timealong an interference path from the transmit antenna to the receiveantenna, such that the radio-frequency corrective signal substantiallycancels self-interference due to signals that propagate along thecorresponding interference path. To perform this cancellation, thecorrective signal is combined with the incoming radio-frequency signalin the receive chain of the transceiver.

In embodiments that make use of multiple parallel cancellation paths, acancellation method involves obtaining a plurality of samples of theoutgoing radio-frequency communication signal and successively adjustingthe phase and magnitude of each of the sampled radio-frequencycommunication signals to generate a plurality of radio-frequencycorrective signals. The plurality of radio-frequency corrective signalswith the incoming radio-frequency signal to reduce the self-interferencecomponent. The parallel cancellation paths may be separated byamplifiers or other forms of radio-frequency isolators. In someembodiments, the parallel cancellation paths are tuned successively,such as through adjustment of parallel delay lines. The delay introducedby each respective delay line preferably corresponds to the travel timealong an interference path from the transmit antenna to the receiveantenna, such that each of the radio-frequency corrective signalssubstantially cancels self-interference due to signal propagation alonga respective interference path.

In some embodiments, the parallel cancellation paths are frequencyselective, with each cancellation path being obtained at a differentfrequency and conveying samples from different portions of the frequencyband. In such embodiments, each of the cancellation paths providesradio-frequency correction in a different segment of the frequency band.This allows separate adjustment of the phase and amplitude of correctivesignals at different frequencies to allow for potentialfrequency-dependent attenuation along various self-interference signalpaths.

Illustrated in FIG. 3 is a method performed at a full-duplex transceiverfor determining nonlinear distortion in a self-interference signal usinga sideband test signal. In step 302, the transceiver generates anoutgoing baseband signal. In addition, in step 304, the transceivergenerates at least one sideband test signal. The sideband test signalhas a center frequency different from the center frequency of theoutgoing baseband signal. In an example that is particularly useful inOFDM systems, in which the baseband test signal includes a plurality ofsubcarrier signals, the sideband test signal also includes a pluralityof subcarrier signals. In some embodiments, two sideband test signalsare generated, one with a center frequency lower than the centerfrequency of the outgoing baseband signal and one with a centerfrequency higher than the center frequency of the outgoing basebandsignal.

In step 306, the test signal is combined with the outgoing basebandsignal. The combination of the test signal with the outgoing basebandsignal may be performed in the frequency domain. For example, a combinedsignal can be generated using a 256-point fast Fourier transform. Insuch an embodiment, the lowest 64 frequencies may represent a first testsignal, followed by 64 frequencies that are set to zero amplitude toprovide a first guard band, followed by 64 frequencies representing theoutgoing baseband signal, then another guard band of 64 frequencies setto zero amplitude. These frequencies may then be converted to a singletime-domain signal by, for example, a dedicated inverse FFT chip.

In step 308, the combined signal is translated, or upconverted, to radiofrequency via a mixing operation. By combining the outgoing basebandsignal and the test signal and then upconverting those signals to radiofrequency, the transceiver generates a combined radio-frequencycommunication signal that includes (i) an outgoing radio-frequencycommunication signal having a center frequency and (ii) aradio-frequency test signal that has a center frequency different fromthe center frequency of the outgoing radio-frequency communicationsignal. Other techniques for generating such a combined radio-frequencycommunication signal could also be employed. For example, the testsignal and the outgoing baseband signal could be separately modulated toradio frequency and then combined. As another example, the combining atthe base-band could be performed using spreading codes, in which casethe test signal and the outgoing base-band signals are spread using twospreading codes and added and as a result these two signals would occupythe same frequency band.

In step 310, the combined radio-frequency signal is amplified with apower amplifier. The amplification with the power amplifier introducesundesirable nonlinear distortions in the amplified signal. For example,the power amplifier may introduce intermodulation distortion among thedifferent frequency components of the amplified signal. In step 312, theamplified signal is transmitted by a transmit antenna of thetransceiver.

In the amplified combined radio frequency signal, the test signalpreferably has a substantially lower power than the communicationsignal. The relative powers of the radio-frequency test signal and theradio-frequency communication signal are chosen such that thecommunication signal is powerful enough to be received by a desiredreceiver, while the test signal is sufficiently weak such that, from theperspective of the desired receiver, the power from the test signalfalls below the level of noise at the desired receiver. To accomplishthis, the radio-frequency test signal may have a power that is, forexample, at least 20 dB less than the power of the radio-frequencycommunication signal. However, the power of test signal is selected tobe significantly above the power of distant signals in the bandsoccupied by the test signal. For example, the radio-frequency testsignal may have a power approximately 40 dB less than the power of theradio-frequency communication signal, and are received (at the sametransmitting node) 20 dB above the received power of distant RF signalsthat may exist in the neighboring frequency bands occupied by the testsignals. In another embodiment, for the purpose of measuring the poweramplifier noise and non-linearity, the power amplifier output is sampledusing a separate receive chain. In such a configuration, to furtherreduce the level of the test signal in the signal received at distantreceivers, filtering can be applied after the component used to samplethe power amplifier output and prior to transmit antenna.

In step 314, the transceiver receives at a receive antenna aradio-frequency signal, which includes a self-interference component dueto the transmission at step 312. Preferably, the transmit antenna andreceive antenna of the transceiver are oriented with respect to oneanother in such a way as to minimize the power of the self-interferencecomponent. For example, the transmit and receive antenna may be pairwisesymmetrical. However, some self-interference from the transmit antennato the receive antenna is unavoidable in practice. To cancel at least aportion of this self-interference component, the transceiver generates aradio-frequency corrective signal in step 316 based on the outgoingbaseband signal. The radio-frequency corrective signal is combined instep 318 so as to cancel at least a portion of the self-interferencecomponent received in step 314.

The radio-frequency corrective signal generated in step 316 is selectedso as to cancel the portion of the self-interference component due tothe transmitted radio-frequency communication signal, but not to cancelthe portion of the self-interference component due to theradio-frequency test signal. This un-cancelled radio-frequency testsignal component will be used to provide a measure of the nonlinearitiesintroduced by the signal amplification in step 310.

In step 320, the radio-frequency corrective signal is demodulated, andin step 322, the demodulated test signal is sampled by one or moreanalog-to-digital converters. Based on the sampled test signal thetransceiver in step 324 determines the nonlinear distortion introducedby the power amplifier. For example, the transceiver may compare thefrequency components of the sampled test signal and compare those withthe original frequency components of the test signal in order todetermine the amount of intermodulation distortion between frequencies.The nonlinear intermodulation distortion between respective frequencycomponents is a function primarily of the differences between thoserespective frequencies, rather than a function of the absolute values ofthe frequencies. Consequently, the intermodulation distortion measuredin the sideband test signal can be used to create a nonlinear model ofthe distortion of the outgoing baseband communication signal. This modelmay be, for example, a model based on a Volterra series, where thekernel coefficients of the Volterra series are measured through thesampling of the sideband test signal.

The transceiver may use the nonlinear model created in step 324 toupdate the model used for creation of the radio-frequency correctivesignal in step 316. For example, the distortion as measured through theuse of the sideband test signal can be applied during the generation ofthe radio frequency corrective signal 316, such that the radio frequencycorrective signal more accurately mimics the self-interference componentand thus leads to more effective signal cancellation in step 318.

The components of exemplary combined radio-frequency communicationsignals are illustrated in FIGS. 4 and 5. In FIG. 4, a combinedradio-frequency communication signal includes a central outgoingradio-frequency signal 400 along with a pair of sideband test signals402 and 404. In the example of FIG. 4, the outgoing radio-frequencysignal consists of several individual subcarrier signals, each with anamplitude dependent on the signal being transmitted. The sideband testsignals 402 and 404 are scaled, frequency-shifted replicas of theoutgoing radio-frequency signal 400. That is, the amplitude of eachsubcarrier signal within the test signals is a scaled-down version ofthe amplitude of a corresponding subcarrier in the outgoingradio-frequency signal. The sideband test signals 402 and 404 arepreferably separated from the outgoing radio-frequency signal 400 byrespective guard bands 406 and 408 in which the transmitted spectralpower is zero (except for power attributable to the intermodulationdistortion itself). The guard bands 406 and 408 help to minimize leakagefrom the outgoing radio frequency signal 400 to the sideband testsignals 402 and 404, and they facilitate separation of the signals. Insuch a configuration, when the test signal is a replica of the outgoingsignal, the non-linearity affects the test signal and the outgoingsignals in similar manners and this feature is used to compensate forthe effect of nonlinearity.

In some embodiments, the sideband test signals need not be scaledreplicas of the outgoing radio-frequency signal. As illustrated in FIG.5, a combined radio-frequency communication signal includes a centraloutgoing radio-frequency signal 500 along with a pair of sideband testsignals 502 and 504. The sideband test signals 502 and 504 are separatedfrom the central outgoing radio-frequency signal 500 by respective guardbands 506 and 508. In this example, the sideband test signals consist ofone or more sparsely-located pilot tones. The pilot tones are locatedamong other frequency slots to which an amplitude of zero is assigned.Intermodulation distortion can be modeled by measuring, in the incomingradio-frequency signal, the effect of the pilot tones on the frequenciesto which zero amplitude was assigned.

As illustrated in FIGS. 4 and 5, the power of the sideband test signalsis lower than the power of the central outgoing radio-frequency signal.It should be noted, however, that the spectral powers illustrated inFIGS. 4 and 5 are not necessarily drawn to scale. Preferably, the levelof sideband test signals are adjusted to have a transmit power that issignificantly lower than the power of the outgoing radio-frequencysignal, and a receive power that is significantly above the receivedpower from the distant RF signals that may exist in those neighboringbands occupied by the test signals. For example, the level of sidebandtest signals are adjusted to have a power that is 40 dB lower than thepower of the central outgoing radio-frequency signal, and are received(at the same node) at a power level which is 20 dB above the receivedpower from the distant RF signals that may exist in those neighboringbands occupied by the test signals. Preferably, the power of thesideband test signals is at least 20 dB greater than the power of adesired radio-frequency signal component.

FIGS. 6 and 7 illustrate exemplary components of a full-duplextransceiver that can be used for sampling an incoming radio-frequencysignal to determine nonlinear distortion. Specifically, the examples ofFIGS. 6 and 7 illustrate components that can be used in the receipt andsampling of signals that include an in-phase (I) component and aquadrature (Q) component. The componentry illustrated in FIGS. 6 and 7allows the use of a single digital-to-analog converter despite thepresence of both I and Q components in the incoming signal, therebysimplifying the componentry to be used.

In FIG. 6, an incoming radio-frequency signal is received at a receiveantenna 600, which may be one of a plurality of receive antennas in aMIMO antenna system. The incoming signal is amplified by a low-noiseamplifier 602. A local oscillator 604 is provided for generating aradio-frequency signal used to demodulate the incoming signal.Preferably, this is the same local oscillator signal used to modulateoutgoing signals of the full-duplex transceiver, which reduces theeffects of jitter on radio frequency interference cancellation. Thesignal from the local oscillator is provided to a phase shifter 606,which splits the local oscillator signal into two separate signals thatdiffer from one another by a phase shift of 90°. The phase shifter 606may be a Schiffman phase shifter. The system includes two frequencymixers 608, 610, each of which is provided with a respective localoscillator signal. The received radio-frequency signal is sent to bothof these mixers 608, 610, one of which extracts the in-phase componentand the other of which extracts the quadrature component of the incomingsignal.

The separate in-phase and quadrature signals are sent through respectivelow-pass filters 612, 614 to remove undesirable high-frequency signalsgenerated in the mixing process. The resulting filtered signals areprovided to switching circuitry 616. Switching circuitry 616 alternatelyswitches between the in-phase and quadrature component to be sampled bya single analog-to-digital converter 618. The analog-to-digitalconverter 618 is synchronized with the switching circuitry 616 such thatthe digital samples taken by the analog-to-digital converter 618alternate repeatedly between a sample of the I component and a sample ofthe Q component. The transceiver is provided with a distortion detectionmodule 622, which quantifies distortion by comparing the signal sampledby the analog-to-digital converter 618 with the outgoing baseband signalfrom the baseband signal generator 620. The distortion detection module622 may operate to generate a nonlinear distortion model by, forexample, calculating the kernel coefficients of a Volterra seriesrepresenting nonlinearity introduced by a power amplifier of thetransceiver.

In an alternative embodiment illustrated in FIG. 7, a radio-frequencysignal is received at the receive antenna 700 and amplified by alow-noise amplifier 702. A local oscillator 704 provides aradio-frequency signal to use in demodulation of the receivedradio-frequency signal. The signal from the local oscillator is providedto a phase shifter 706, which splits the local oscillator signal intotwo separate signals that differ from one another by a phase shift of90°. The phase shifter 706 may be a Schiffman phase shifter. Theseparate outputs of the phase shifter 706 are provided to a switch 716,which switches alternately between the two phase-shifted localoscillator signals. This alternating-phase signal is supplied to a mixer710, which mixes the alternating-phase local-oscillator signal with theamplified incoming signal. Consequently, the output of the mixer 710switches alternately between the demodulated in-phase component I andthe demodulated quadrature component Q. The mixed signal is passedthrough a low-pass filter 712 to reject undesirable high-frequencycomponents. A single analog-to-digital converter 718 is provided toalternately sample the demodulated I and Q components. Theanalog-to-digital converter 718 is synchronized with the switch 716 suchthat the digital samples taken by the analog-to-digital converter 718alternate repeatedly between a sample of the I component and a sample ofthe Q component.

The transceiver is provided with a distortion detection module 722,which quantifies distortion by comparing the signal sampled by theanalog-to-digital converter 718 with the outgoing baseband signal fromthe baseband signal generator 720. The distortion detection module 722may operate to generate a nonlinear distortion model by, for example,calculating the kernel coefficients of a Volterra series representingnonlinearity introduced by a power amplifier of the transceiver.

In the systems of FIGS. 6 and 7, the alternate samples of the I and Qcomponents can be reconstructed by the distortion detection modules 622,722 into a single signal using the following technique. Where each pairof in-phase and quadrature samples is designated as X_(i) and Y_(i),respectively, the combined signal can be treated as a signal withamplitude r_(i) and phase θ_(i), where r_(i)=(X_(i)+Y_(i))^(1/2), andθ_(i)=arctan (X_(i)/Y_(i)).

In some embodiments, the path for the sampling of the power amplifieroutput is designed solely for this purpose. In such cases, the receivedbase-band signals at 618 and 718, in addition to capturing thenon-linearity, provide information about the noise generated by thepower amplifier. It is also possible to have two complete receive chains(with two receive antennas) wherein the tasks of signal reception andpower amplifier sampling are combined. Such an embodiment is illustratedschematically in FIG. 11. In this case, the signals at the base-bands ofthe two received chains contain a combination of the desired signal fromdistant transmitter and a sample of the power amplifier output(self-interference). These two signals are combined to maximize thesignal-to-noise ratio of the desired signal from distant transmitter.The use of multiple antennas may then serve the dual role of not onlyhelping with the cancellation of self-interference, but also with theincrease in receive diversity. Mathematical expressions to maximize thesignal-to-noise ratio of the desired signal from distant transmitterprovide the best compromise between these two objectives. Note that thisis different from standard maximum ratio combining used in multiplereceive antennas. In exemplary embodiments, it is not a requirement forthe receiver to recover the self-interference; rather, the goal issimply to minimize the impact of self-interference towards maximizingthe signal-to-noise ratio of the signal from distant (desired)transmitter. Note that the number of transmit antennas can be alsoincreased to two, wherein one transmit antenna is used to transmit theRF corrective signal. In another embodiment, a 2×2 antenna transceiveris configured to function in the following modes of operations dependingon the rate requirements: (1) a 2×2 half-duplex mode, (2) a 1×1full-duplex mode, (3) a combination of modes (1) and (2) wherein someOFDM tones send two streams of data from A to B, while the rest of thetones send one stream from A to B and one stream from B to A.

Another embodiment concerns a dual mode full-duplex node with a MIMOmode and a full-duplex mode. Referring to FIG. 11, there is anadditional receive chain (middle chain) used to sample the PA output. InFIG. 11, this chain is connected to a second receive antenna and the twobase-band signals are combined to cancel the self-interference (againaccounting for PA non-linearity and noise), while achieving receivediversity. Combining is achieved using a generalized version of maximumratio combining.

FIGS. 12 to 16 relate to configurations that the full-duplex nodes aredesigned to operate in two modes, namely a full-duplex mode and a MIMOmode (where both antennas send data in one direction). The aim is tominimize the number of TX/RX chains such that both these configurationsare supported. Although here the concept is explained in terms of 2×2systems, similar arguments are applicable to systems with more TX and/orRX chains.

FIGS. 12 and 13 show the case of a legacy TDD MIMO where the chains infull line are active and those in dashed lines are idle, depending onthe units being in TX (FIG. 12) or RX (FIG. 13) modes. Here, we aim touse the idle chains to play the roles of the auxiliary chain forcorrective signal injection, and the chain for sampling PA output. FIGS.14 to 16 correspond to different configurations achieving this goal.

FIG. 14 shows one possible configuration in which the output of PA issampled through the circulator (which provides about 30 dB isolation).This reduces the complexity on the conversion between an actual RX chainto a chain used to sample the PA output. However, the PA in the TX chainbuilding the corrective signal is bypassed to avoid its noise, seeswitches S₁-S₁′ and S₂-S₂′, where positions S₁ and S₂ put the chain inactual TX mode (MIMO mode), while positions S₁′ and S₂′ put the chain inthe mode of forming the corrective signal (full-duplex mode).

FIG. 15 shows a configuration with a higher complexity on which bothchains are adjusted to operate in MIMO or full-duplex mode. See switchesS₁-S₁′, S₂-S₂′, S₃-S₃′, S₄-S₄′ where positions S₁, S₂, S₃, S₄ put thechains in actual TX mode (MIMO mode), while positions S₁′, S₂′, S₃′, S₄′put the chains in the full-duplex mode. Switches can be pin diode,relays or MEMS.

FIG. 16 shows a configuration similar to the one in FIG. 15, but with adifferent approach to bypassing the PA (see switches S₃-S₃′ and S₄-S₄′),which allows for a higher speed of operation as the current in the PAcontinues flowing.

FIGS. 15 and 16 are two extreme cases for the purpose of putting the PAinto sleep to avoid its output noise in the corrective chain, andintermediate options such as reducing voltage and current of PA are alsopossible.

In the embodiments explained in FIGS. 14, 15, 16, we use an auxiliarytransmit (ATX) chain to generate the corrective signal, and an auxiliaryreceive (ARX) chain to sample the PA output. FIG. 17 shows a low costimplementation for such setup. Although the locations of the terminalare shown to be centered and symmetrical, the actual locations areadjusted to provide the desired coupling depending on the levels of thedifferent signals. Another embodiment includes two such patches, onewith terminals PTX (primary TX), and ARX, and the other one withterminals for primary RX (PRX) and ATX. Although the concept isexplained in terms of patch antennas, other forms of multi-terminalantennas are possible.

To further reduce the hardware complexity, FIG. 18 shows an embodimentusing an auxiliary corrective signal for analog active cancellation inanalog base-band, and an auxiliary base-band receive signal usedsampling of the PA output to capture PA non-linearity effect as well asPA noise. The two analog base-band signals at points 1801 and 1802contain the leaked signal, including the effects of PA non-linearity andnoise. Analog active cancellation is performed at the base-band using anOperation Amplifier (OP-Amp). Relative signal gains and base-band analogcancellation are adjusted such that the signal at point 1802 isdominated by the leaked signal (residual self-interference), while thecomponent of the leaked signal at point 1801 is kept as low as possible.Signal at point 1802 contains the leaked signal with a highSignal-to-Noise ratio, which upon conversion to digital domain providesa sample of the PA output including the effects of PA non-linearity andnoise to be used for the cancellation of the copy of the same signalleaked into the base-band of the received signal.

To reduce the required dynamic range of the LNA, FIGS. 19 to 22 showembodiments using two auxiliary corrective signals, one used for analogactive cancellation at low frequency (analog base-band, or analog IF)and another one for analog active cancellation in the RF domain prior toLNA. Similar to the embodiment in FIG. 18, an auxiliary base-bandreceive signal is used for sampling of the PA output to capture PAnon-linearity effect as well as PA noise. The two analog base-bandsignals at points 1901 and 1902 contain the leaked signal, including theeffects of PA non-linearity and noise. Base-band (or IF) analog activecancellation is performed at the base-band (or IF) using an OperationAmplifier (OP-Amp), or other known hardware for adding analog signals.Relative signal gains, and RF as well as BB (or IF) analog cancellation,are adjusted such that the signal at point 1902 is dominated by theleaked signal (residual self-interference), while the component of theleaked signal at point 1901 is kept as low as possible. As a result,signal at point 1902 contains the leaked signal with a highSignal-to-Noise ratio, which upon conversion to digital domain providesa sample of the PA output including the effects of PA non-linearity andnoise to be used for the cancellation of the copy of the same signalleaked into the base-band of the received signal. FIGS. 20 to 22 arehardware simplifications of the basic method shown in FIG. 19, whereinthe two corrective signals are formed jointly, and are frequencymultiplexed.

Amplify and Forward (AF) relaying concerns setups where the relay simplyamplifies its received signal and forwards it. In AF relaying, if thedelay introduced by any given relay is less than the cyclic prefix ofthe underlying OFDM, the combined signal received at a destination willmaintain its OFDM structure (FIGS. 23, 24, 25). This is a desirablefeature, as in this case the inclusion of AF relays will be transparentto the end nodes. However, as the transmitter and the receiver of the AFrelays operate over the same time and frequency, the issue ofself-interference needs to be addressed. Methods described hereinprovide the tools to reduce the amount of self-interference to anacceptable level. However, to avoid interruption in the operation of AFrelay, the issue of training is addressed. An embodiment superimposes alow power training signal, after spreading with a long pseudo-randomsequence, on top of the amplify/forward signal. Receiver of the AFrelay, after dispreading the training sequence and realizing theembedded processing gain, will use the result to estimate the channelfor the self-interference. Two such training signals are needed tomeasure the channels corresponding to the residual self-interferencefrom the primary transmit chain to the receive base-band, and thechannel from the corrective signal path to the receive base-band. Thesetwo signals can be multiplexed in time, or sent simultaneously using twodifferent spreading codes. In addition, this embodiment includes the useof adaptive methods to update the estimates of these two channels. Thisoperation, which essentially entails measuring the two channels' impulseresponses, is similar to the adaptive equalization techniques reportedin the literature, which can be performed blindly (no or little trainingis sent). This operation is repeated periodically to update themeasurement of the self-interference channel Processing gain is selectedto be high enough such that the interference caused at the finaldestination (due to the superimposed training signal) is negligible(i.e., falls below the noise level). FIG. 26 shows another embodiment,wherein, to reduce hardware complexity, filtering for the constructionof the corrective signal is applied directly to transmit base-bandsignal. Legacy wireless standards rely on either Frequency DivisionDuplex (FDD) or Time Division Duplex (TDD) to separate the uplink andthe down-link. It is desirable that the operation of the full-duplex AFrelay remains transparent to legacy nodes. This requires adjusting thedirection of the relaying in TDD systems. Methods of this embodimentrely on detecting the preambles corresponding to the uplink anddown-link to adjust the direction of AR relaying accordingly. This isachieved by searching for the preamble in the receive chain of the TDDsystem. Upon detecting the preamble, a decision is made to eitherreverse the direction of AF relaying, or not, depending on the level andtype of the detected preamble and its relative time of occurrence.Reversing the direction of AF relaying can be accompanied by exchangingthe roles of the TX and RX antennas, such that the polarization of eachtransmit antenna is matched to the polarization of its correspondingreceiving antenna, and vice versa. Normally, the down-conversion is fromRF to base-band, which would necessitate the use of two A/D and two D/Afor each channel of I/Q. In a system with an auxiliary transmit andauxiliary receive units, this results in a total of 4 A/Ds and 4 D/As,which may be costly for some applications. A solution to this problem isthe use of an Intermediate Frequency (IF) close to the base-band, inwhich case the second stage of down-conversion from the IF to the actualbase-band is performed in the digital domain. In such a case, a singleA/D and a single D/A will be sufficient for each receive and transmitchains, respectively, resulting in a total of two A/D and two D/A for asystem with an auxiliary transmitter and an auxiliary receiver.

The modular components described herein, such as the digital decodingmodule 148, the baseband correction module 106, the radio-frequencycorrection module 120, and the distortion detection modules 622, 722,may be implemented by general purpose processors or by programmablelogic devices programmed to perform the functions described herein.Alternatively, those components may be implemented with the use ofspecial-purpose digital circuitry.

The systems and methods described above are intended as examples, andthe scope of the invention is not limited to those examples. Rather, thescope of the invention is delineated by the following claims.

1. A method comprising: filtering an outgoing signal via an adaptivecorrective filter at a transmitter; transmitting the outgoing signalwhich causes self-interference to a receiver coupled to the transmitter,the receiver and the transmitter configured for cable modemapplications, the filtered signal providing a corrective signal forcorrecting self-interference; receiving an incoming data signal at thereceiver; and adding the corrective signal to the received data signalto reduce self-interference from the transmitter to the receiver.
 2. Themethod of claim 1 wherein the transmitter and the receiver collectivelyform a full-duplex data transceiver.
 3. The method of claim 1 whereinthe transmitter and the receiver are configured as a full-duplexamplify-and-forward relay to assist a central node in serving one ormore distant client nodes.
 4. The method of claim 1 further comprising:down-converting the received data signal, wherein the corrective signalis added to the received data signal at a first correction point along areceive chain prior to down-converting the received data signal.
 5. Themethod of claim 4 further comprising: adding a second corrective signalat a second correction point along the receive chain after thedown-conversion of the received data signal.
 6. The method of claim 5,wherein the adaptive corrective filter is one of a plurality of adaptivecorrective filters, each adaptive corrective filter trained by:transmitting a dedicated training signal, each dedicated training signalconfigured to be detected at the receiver to enable the receiver tomeasure a channel corresponding to a respective residualself-interference correction term; and applying the measurement of eachsuch channel to adapt its respective adaptive corrective filter toenable cancellation of the respective residual self-interference at thefirst correction point or the second correction point.
 7. The method ofclaim 6 wherein each dedicated training signal is one or more of timedomain multiplexed before transmission in an outgoing signal, frequencydomain multiplexed before transmission in an outgoing signal, or codedivision multiplexed before transmission in an outgoing signal.
 8. Themethod of claim 7 further comprising: measuring one or morenonlinearities via one or more of the dedicated training signals; usingthe measured one or more nonlinearities to adjust the correction of theresidual self-interference.
 9. The method of claim 8 further comprising:adding a third corrective signal at a third correction point along thereceive chain after the down-conversion of the received data signal. 10.The method of claim 9, wherein the adaptive corrective filter is one ofthe plurality of adaptive corrective filters, each adaptive correctivefilter trained by: transmitting a dedicated training signal, eachdedicated training signal configured to be detected at the receiver toenable the receiver to measure a channel corresponding to a respectiveresidual self-interference correction term; and applying the measurementof each such channel to adapt each respective adaptive corrective filterto enable cancellation of the respective residual self-interference atone or more of the first correction point, the second correction pointor the third correction point.
 11. The methods of claim 10, wherein thetransmitter and the receiver are configured as a full-duplex datatransceiver configured for cable modem applications, the full-duplextransceiver configured to multiplex a plurality of upstream signals anda plurality of downstream signals, the full-duplex transceiverconfigured to exchange data using upstream and downstream signals over anetwork, the network including a plurality of full-duplex client nodesand a plurality of half-duplex client nodes.
 12. The method of claim 11further comprising: broadcasting the plurality of downstream signalsfrom the full-duplex data transceiver configured as a central node, theplurality of downstream signals being addressed to one or more of theclient nodes; the central node, upon request by one or more of theclient nodes to transmit upstream data, allocates at least onedesignated client node of the plurality of client nodes to send upstreamdata to the central node; and receiving the plurality of upstreamsignals at the central node.
 13. The method of claim 12 wherein thefull-duplex central node, while receiving data in the plurality ofupstream signals from one of the plurality of client nodes, transmits acentral node jamming signal in the same band to jam signals for one ormore unwanted receivers in case the plurality of upstream signals iseavesdropped, and wherein one of the plurality of full-duplex clientnodes, while receiving data in the plurality of downstream signals fromthe central node, transmits a client jamming signal in the same band tojam signals for one or more unwanted receivers in case the plurality ofdownstream signals is eavesdropped.
 14. A shared cable network toprovide high-speed access to packet-based data services, the sharedcable network comprising: a full-duplex central node; a plurality clientnodes coupled to the full-duplex central node, the plurality of clientnodes including half-duplex and full-duplex nodes, wherein thefull-duplex central node is configured to: filter downstreampacket-based data via an adaptive corrective filter; broadcast thedownstream packet-based data to the plurality of client nodes, thebroadcast packet-based data causing self-interference to a receiverwithin the full-duplex central node, the filtered broadcast downstreampacket-based data providing a corrective signal for correctingself-interference; receive upstream packet-based data at the receiver;and add the corrective signal to the received upstream packet-based datato reduce self-interference at the receiver, the received upstreampacket-based data from an allocated full-duplex client node identifiedas a dual-purpose client node to enable receiving the upstreampacket-based data from the dual-purpose client node while thedual-purpose client node receives the broadcast data from thefull-duplex central node.
 15. The shared cable network of claim 14wherein the full-duplex central node is configured to multiplex thebroadcast downstream packet-based data and the upstream packet-baseddata in one or more of a frequency domain or a time domain.
 16. Theshared cable network of claim 15 wherein the full-duplex central node isconfigured to multiplex the broadcast downstream packet-based data andthe upstream packet-based data with frequency division multiplexing,wherein the allocated dual-purpose client node receives downstreampacket-based data from the full-duplex central node in an upstreamfrequency band and during a time slot allocated by the full-duplexcentral node, while simultaneously sending upstream packet-based data tothe full-duplex central node in the same upstream frequency band andduring the same time slot, and receives downstream packet-based datafrom the full-duplex central node in a downstream frequency band whilesimultaneously sending upstream data to the full-duplex central node inthe same downstream frequency band upon adhering to a wait time to avoidcausing interference to the other client nodes that are still listeningto a downstream signal in order to identify the recipient of downstreampackets and extract common data broadcasted to the plurality of clientnodes.
 17. The shared cable network of claim 15 wherein the full-duplexcentral node is configured to multiplex the broadcast downstreampacket-based data and the upstream packet-based data with time-divisionmultiplexing, wherein the allocated dual-purpose client node receivesdownstream packet-based data from the full-duplex central node during anupstream time slot allocated by the full-duplex central node whilesimultaneously sending upstream packet-based data to the full-duplexcentral node during the same upstream time slot, and receives downstreampacket-based data from the full-duplex central node during thedownstream time slot while simultaneously sending upstream data to thefull-duplex central node during the same downstream time slot uponadhering to a wait time to avoid causing interference to the otherclient nodes that are still listening to the downstream signal in orderto identify a target recipient of downstream packets and extract commondata broadcasted to the plurality of client nodes.
 18. The shared cablenetwork of claim 15 with frequency division multiplexing whereindownstream data is broadcasted by the full-duplex central node in adownstream frequency band to the network of half-duplex and full-duplexclient nodes while an upstream frequency band is allocated by thefull-duplex central node to one of the plurality of half-duplex orfull-duplex client nodes, and wherein, if the client node allocated bythe central node to transmit in the upstream band has full-duplexcapability, the said client node is entitled to simultaneously receivedownstream data from the full-duplex central node in the same upstreamfrequency band, and wherein, if the client node addressed by the centralnode to receive and decode the downstream data packets has full-duplexcapability, the said client node is entitled to simultaneously sendupstream data packets to the full-duplex central node in the samedownstream frequency band upon adhering to a wait time to avoid causinginterference to the other client nodes that are still listening to thedownstream signal in order to identify a target recipient of downstreampackets and extract common data broadcasted to the plurality of clientnodes.
 19. The shared cable network of claim 15 with time divisionmultiplexing wherein downstream data is broadcasted by the full-duplexcentral node in a downstream time slot to the network of half-duplex andfull-duplex client nodes while an upstream time slot is allocated by thefull-duplex central node to one of the plurality of half-duplex orfull-duplex client nodes, and wherein, if the client node allocated bythe central node to transmit in the upstream time slot has full-duplexcapability, the said client node is entitled to simultaneously receivedownstream data from the full-duplex central node in the same upstreamtime slot, and wherein, if the client node addressed by the central nodeto receive and decode the downstream data packets has full-duplexcapability, the said client node is entitled to simultaneously sendupstream data to the full-duplex central node in the same downstreamtime slot upon adhering to a wait time to avoid causing interference tothe other client nodes that are still listening to a downstream signalin order to identify a target recipient of downstream packets andextract common data broadcasted to the plurality of client nodes.